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1 lt1229/lt1230 typical applicatio u applicatio s u descriptio u features dual and quad 100mhz current feedback amplifiers the lt ? 1229/lt1230 dual and quad 100mhz current feedback amplifiers are designed for maximum perfor- mance in small packages. using industry standard pinouts, the dual is available in the 8-pin minidip and the 8-pin so package while the quad is in the 14-pin dip and 14-pin so. the amplifiers are designed to operate on almost any available supply voltage from 4v ( 2v) to 30v ( 15v). these current feedback amplifiers have very high input impedance and make excellent buffer amplifiers. they maintain their wide bandwidth for almost all closed-loop voltage gains. the amplifiers drive over 30ma of output current and are optimized to drive low impedance loads, such as cables, with excellent linearity at high frequencies. the lt1229/lt1230 are manufactured on linear technologys proprietary complementary bipolar process. for a single amplifier like these see the lt1227 and for better dc accuracy see the lt1223. n 100mhz bandwidth n 1000v/ m s slew rate n low cost n 30ma output drive current n 0.04% differential gain n 0.1 differential phase n high input impedance: 25m w , 3pf n wide supply range: 2v to 15v n low supply current: 6ma per amplifier n inputs common mode to within 1.5v of supplies n outputs swing within 0.8v of supplies n video instrumentation amplifiers n cable drivers n rgb amplifiers n test equipment amplifiers + v out lt1229 ?ta01 12.1k r f2 750 w 1% resistors worst case cmrr = 22db typically = 38db v out = g (v in + ?v in ) r f1 = r f2 r g1 = (g ?1) r f2 r g2 = trim cmrr with r g1 high input resistance does not load cable even when power is off 1/2 lt1229 r f2 g ?1 r g2 187 w r f1 750 w r g1 3.01k + 1/2 lt1229 3.01k 3.01k 12.1k v in v in + bnc inputs video loop through amplifier loop through amplifier frequency response frequency (hz) 10 ?0 gain (db) ?0 ?0 ?0 ?0 ?0 10 100 1k 10k 100m lt1229 ?ta02 100k 1m 10m 0 common mode signal normal signal , ltc and lt are registered trademarks of linear technology corporation.
lt1229/lt1230 2 a u g w a w u w a r b s o lu t exi t i s supply voltage ...................................................... 18v input current ...................................................... 15ma output short circuit duration (note 2) ......... continuous operating temperature range lt1229c, lt1230c ............................... 0 c to 70 c lt1229m, lt1230m (obsolete).. C55 c to 125 c storage temperature range ..................C65 c to 150 c junction temperature plastic package .............................................. 150 c ceramic package (obsolete) ................ 175 c lead temperature (soldering, 10 sec.)................. 300 c order part number lt1230cn lt1230cs order part number s8 part marking lt1229cn8 lt1229cs8 1229 wu u package / o rder i for atio s package 14-lead plastic soic n package 14-lead plastic dip + v d 14 13 12 11 10 9 8 7 6 5 4 3 2 1 out a ?n a +in a +in b ?n b out b out c v ?n d out d top view a +in d +in c ?n c c b 8 7 6 5 4 3 2 1 + ?n a +in a v top view n8 package 8-lead plastic dip out a out b v ?n b +in b a b s8 package 8-lead plastic soic t j max = 150 c, q ja = 100 c/w (n8) t j max = 150 c, q ja = 150 c/w (s8) (note 1) t j max = 175 c, q ja = 100 c/w (j8) obsolete package consult ltc marketing for parts specified with wider operating temperature ranges. lt1229mj8 lt1229cj8 consider the n package for alternate source lt1230mj lt1230cj t j max = 150 c, q ja = 70 c/w (n) t j max = 150 c, q ja = 110 c/w (s) obsolete package consider the n package for alternate source order part number j8 package 8-lead ceramic dip t j max = 175 c, q ja = 80 c/w (j) j package 14-lead ceramic dip order part number
3 lt1229/lt1230 symbol parameter conditions min typ max units v os input offset voltage t a = 25 c 3 10 mv l 15 mv input offset voltage drift l 10 m v/ c i in + noninverting input current t a = 25 c 0.3 3 m a l 10 m a i in C inverting input current t a = 25 c 10 50 m a l 100 m a e n input noise voltage density f = 1khz, r f = 1k, r g = 10 w , r s = 0 w 3.2 nv/ ? hz +i n noninverting input noise current density f = 1khz, r f = 1k, r g = 10 w , r s = 10k 1.4 pa/ ? hz Cin inverting input noise current density f = 1khz 32 pa/ ? hz r in input resistance v in = 13v, v s = 15v l 225 m w v in = 3v, v s = 5v l 225 m w c in input capacitance 3pf input voltage range v s = 15v, t a = 25 c 13 13.5 v l 12 v v s = 5v, t a = 25 c 3 3.5 v l 2v cmrr common mode rejection ratio v s = 15v, v cm = 13v, t a = 25 c5569db v s = 15v, v cm = 12v l 55 db v s = 5v, v cm = 3v, t a = 25 c5569db v s = 5v, v cm = 2v l 55 db inverting input current v s = 15v, v cm = 13v, t a = 25 c 2.5 10 m a/v common mode rejection v s = 15v, v cm = 12v l 10 m a/v v s = 5v, v cm = 3v, t a = 25 c 2.5 10 m a/v v s = 5v, v cm = 2v l 10 m a/v psrr power supply rejection ratio v s = 2v to 15v, t a = 25 c6080db v s = 3v to 15v l 60 db noninverting input current v s = 2v to 15v, t a = 25 c1050na/v power supply rejection v s = 3v to 15v l 50 na/v inverting input current v s = 2v to 15v, t a = 25 c 0.1 5 m a/v power supply rejection v s = 3v to 15v l 5 m a/v e lectr ic al c c hara terist ics the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. each amplifier, v cm = 0v, 5v v s = 15v, pulse tested unless otherwise noted.
lt1229/lt1230 4 the l denotes the specifications which apply over the full operating temperature range, otherwise specifications are at t a = 25 c. each amplifier, v cm = 0v, 5v v s = 15v, pulse tested unless otherwise noted. symbol parameter conditions min typ max units a v large-signal voltage gain, (note 3) v s = 15v, v out = 10v, r l = 1k l 55 65 db v s = 5v, v out = 2v, r l = 150 w l 55 65 db r ol transresistance, d v out / d i inC , (note 3) v s = 15v, v out = 10v, r l = 1k l 100 200 k w v s = 5v, v out = 2v, r l = 150 w l 100 200 k w v out maximum output voltage swing, (note 3) v s = 15v, r l = 400 w , t a = 25 c 12 13.5 v l 10 v v s = 5v, r l = 150 w , t a = 25 c 3 3.7 v l 2.5 v i out maximum output current r l = 0 w , t a = 25 c 30 65 125 ma i s supply current, (note 4) v out = 0v, each amplifier, t a = 25 c 6 9.5 ma l 11 ma sr slew rate, (notes 5 and 7) t a = 25 c 300 700 v/ m s sr slew rate v s = 15v, r f = 750 w , r g = 750 w , r l = 400 w 2500 v/ m s t r rise time, (notes 6 and 7) t a = 25 c1020ns bw small-signal bandwidth v s = 15v, r f = 750 w , r g = 750 w , r l = 100 w 100 mhz t r small-signal rise time v s = 15v, r f = 750 w , r g = 750 w , r l = 100 w 3.5 ns propagation delay v s = 15v, r f = 750 w , r g = 750 w , r l = 100 w 3.5 ns small-signal overshoot v s = 15v, r f = 750 w , r g = 750 w , r l = 100 w 15 % t s settling time 0.1%, v out = 10v, r f =1k, r g = 1k, r l =1k 45 ns differential gain, (note 8) v s = 15v, r f = 750 w , r g = 750 w , r l = 1k 0.01 % differential phase, (note 8) v s = 15v, r f = 750 w , r g = 750 w , r l = 1k 0.01 deg differential gain, (note 8) v s = 15v, r f = 750 w , r g = 750 w , r l = 150 w 0.04 % differential phase, (note 8) v s = 15v, r f = 750 w , r g = 750 w , r l = 150 w 0.1 deg note 1: absolute maximum ratings are those values beyond which the life of a device may be impaired. note 2: a heat sink may be required depending on the power supply voltage and how many amplifiers are shorted. note 3: the power tests done on 15v supplies are done on only one amplifier at a time to prevent excessive junction temperatures when testing at maximum operating temperature. note 4: the supply current of the lt1229/lt1230 has a negative temperature coefficient. for more information see the application information section. note 5: slew rate is measured at 5v on a 10v output signal while operating on 15v supplies with r f = 1k, r g = 110 w and r l = 400 w . the slew rate is much higher when the input is overdriven and when the amplifier is operated inverting, see the applications section. note 6: rise time is measured from 10% to 90% on a 500mv output signal while operating on 15v supplies with r f = 1k, r g = 110 w and r l = 100 w . this condition is not the fastest possible, however, it does guarantee the internal capacitances are correct and it makes automatic testing practical. note 7: ac parameters are 100% tested on the ceramic and plastic dip packaged parts (j and n suffix) and are sample tested on every lot of the so packaged parts (s suffix). note 8: ntsc composite video with an output level of 2v p . e lectr ic al c c hara terist ics
5 lt1229/lt1230 cc hara terist ics uw a t y p i ca lper f o r c e voltage gain and phase vs C 3db bandwidth vs supply C 3db bandwidth vs supply frequency, gain = 40db voltage, gain = 100, r l = 100 w voltage, gain = 100, r l = 1k w voltage gain and phase vs C 3db bandwidth vs supply C 3db bandwidth vs supply frequency, gain = 20db voltage, gain = 10, r l = 100 w voltage, gain = 10, r l = 1k voltage gain and phase vs C 3db bandwidth vs supply C 3db bandwidth vs supply frequency, gain = 6db voltage, gain = 2, r l = 100 w voltage, gain = 2, r l = 1k supply voltage ( v) 2 ?db bandwidth (mhz) 40 100 120 12 16 lt1229 ?tpc05 4 068101418 0 20 60 140 160 180 r f = 500 w 80 peaking 0.5db peaking 5db r f = 750 w r f = 1k r f = 2k r f = 250 w supply voltage ( v) 2 ?db bandwidth (mhz) 40 100 120 12 16 lt1229 ?tpc06 4 068101418 0 20 60 140 160 180 r f = 500 w 80 peaking 0.5db peaking 5db r f = 750 w r f = 1k r f = 2k r f = 250 w supply voltage ( v) 2 ?db bandwidth (mhz) 4 10 12 12 16 lt1229 ?tpc08 4 068101418 0 2 6 14 16 18 r f = 500 w 8 r f = 1k r f = 2k supply voltage ( v) 2 ?db bandwidth (mhz) 4 10 12 12 16 lt1229 ?tpc09 4 068101418 0 2 6 14 16 18 r f = 500 w 8 r f = 1k r f = 2k frequency (mhz) 0 voltage gain (db) 2 4 6 8 0.1 10 100 lt1229 ?tpc01 ? 1 7 5 3 1 ? phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 w r f = 750 w supply voltage ( v) 2 ?db bandwidth (mhz) 40 100 120 12 16 lt1229 ?tpc02 4 068101418 0 20 60 140 160 180 r f = 500 w 80 peaking 0.5db peaking 5db r f = 750 w r f = 1k r f = 2k supply voltage ( v) 2 ?db bandwidth (mhz) 40 100 120 12 16 lt1229 ?tpc03 4 068101418 0 20 60 140 160 180 80 peaking 0.5db peaking 5db r f = 750 w r f = 1k r f = 2k r f = 500 w frequency (mhz) 14 voltage gain (db) 16 18 20 22 0.1 10 100 lt1229 ?tpc04 12 1 21 19 17 15 13 phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 w r f = 750 w frequency (mhz) 34 voltage gain (db) 36 38 40 42 0.1 10 100 lt1229 ?tpc07 32 1 41 39 37 35 33 phase shift (deg) 180 90 0 45 135 225 phase gain v s = 15v r l = 100 w r f = 750 w
lt1229/lt1230 6 cc hara terist ics uw a t y p i ca lper f o r c e input common mode limit vs output saturation voltage vs output short-circuit current vs temperature temperature junction temperature maximum capacitance load vs total harmonic distortion vs 2nd and 3rd harmonic feedback resistor frequency distortion vs frequency frequency (hz) total harmonic distortion (%) 0.01 0.10 10 1k 10k 100k lt1229 ?tpc11 0.001 100 v s = 15v r l = 400 w r f = r g = 750 w v o = 7v rms v o = 1v rms temperature ( c) ?5 output short circuit current (ma) 40 60 100 150 lt1229 ?tpc15 0 50 25 50 75 125 175 30 70 50 frequency (hz) 10 1 10 100 1k 100k lt1229 ?tpc16 100 10k spot noise (nv/ ? hz or pa/ ? hz) ? n e n +i n frequency (hz) output impedance ( w ) 0.1 100 10k 1m 10m 100m lt1229 ?tpc18 0.001 100k 0.01 10 v s = 15v 1.0 r f = r g = 2k r f = r g = 750 w feedback resistor (k w ) 10 capacitive load (pf) 100 1000 10000 023 lt1229 ?tpc10 1 1 v s = 5v v s = 15v r l = 1k peaking 5db gain = 2 frequency (mhz) 1 ?0 distortion (dbc) ?0 ?0 ?0 ?0 ?0 10 100 lt1229 ?tpc12 v s = 15v v o = 2v p-p r l = 100 w r f = 750 w a v = 10db 2nd 3rd temperature ( c) common mode range (v) 2.0 v + 50 25 75 125 lt1229 ?tpc13 v 0 1.0 ?.0 ?.0 0.5 ?.5 1.5 0.5 25 50 100 v + = 2v to 18v v = ?v to ?8v temperature ( c) output saturation voltage (v) v + 50 25 75 125 lt1229 ?tpc14 v 0 1.0 ?.0 0.5 0.5 ?5 50 100 r l = 2v v s 18v spot noise voltage and current vs power supply rejection vs output impedance vs frequency frequency frequency frequency (hz) power supply rejection (db) 40 80 10k 1m 10m 100m lt1229 ?tpc17 0 100k v s = 15v r l = 100 w r f = r g = 750 w negative 20 60 positive
7 lt1229/lt1230 settling time to 10mv vs settling time to 1mv vs output step output step supply current vs supply voltage w i spl ii f ed s w a ch e ti c cc hara terist ics uw a t y p i ca lper f o r c e one amplifier supply voltage ( v) supply current (ma) 12 lt1229 ?tpc21 4 0816 0 10 5 1 2 3 4 6 7 8 9 2 6 10 14 18 ?5 c 25 c 125 c 175 c settling time (ns) output step (v) 60 lt1229 ?tpc19 20 0 40 80 100 ?0 10 0 ? ? ? ? 2 4 6 8 noninverting inverting v s = 15v r f = r g = 1k inverting noninverting settling time ( m s) output step (v) 12 lt1229 ?tpc20 4 0 8 16 20 ?0 10 0 ? ? ? ? 2 4 6 8 noninverting inverting v s = 15v r f = r g = 1k noninverting inverting lt1229 ?ta03 +in ?n v out v + v
lt1229/lt1230 8 limited by the gain bandwidth product of about 1ghz. the curves show that the bandwidth at a closed-loop gain of 100 is 10mhz, only one tenth what it is at a gain of two. capacitance on the inverting input current feedback amplifiers want resistive feedback from the output to the inverting input for stable operation. take care to minimize the stray capacitance between the output and the inverting input. capacitance on the inverting input to ground will cause peaking in the frequency response (and overshoot in the transient response), but it does not degrade the stability of the amplifier. the amount of capacitance that is necessary to cause peaking is a func- tion of the closed-loop gain taken. the higher the gain, the more capacitance is required to cause peaking. we can add capacitance from the inverting input to ground to increase the bandwidth in high gain applications. for example, in this gain of 100 application, the bandwidth can be increased from 10mhz to 17mhz by adding a 2200pf capacitor. lt1229 ?ta05 + c g r g 5.1 w r f 510 w v out 1/2 lt1229 v in boosting bandwidth of high gain amplifier with capacitance on inverting input frequency (mhz) 1 19 gain (db) 22 25 28 31 46 49 10 100 lt1229 ?ta06 34 37 40 43 c g = 4700pf c g = 2200pf c g = 0 u s a o pp l ic at i wu u i for atio the lt1229/lt1230 are very fast dual and quad current feedback amplifiers. because they are current feedback amplifiers, they maintain their wide bandwidth over a wide range of voltage gains. these amplifiers are designed to drive low impedance loads such as cables with excellent linearity at high frequencies. feedback resistor selection the small-signal bandwidth of the lt1229/lt1230 is set by the external feedback resistors and the internal junction capacitors. as a result, the bandwidth is a function of the supply voltage, the value of the feedback resistor, the closed-loop gain and load resistor. the characteristic curves of bandwidth versus supply voltage are done with a heavy load (100 w ) and a light load (1k) to show the effect of loading. these graphs also show the family of curves that result from various values of the feedback resistor. these curves use a solid line when the response has less than 0.5db of peaking and a dashed line when the re- sponse has 0.5db to 5db of peaking. the curves stop where the response has more than 5db of peaking. small-signal rise time with r f = r g = 750 w , v s = 15v, and r l = 100 w lt1229 ? ta04 at a gain of two, on 15v supplies with a 750 w feedback resistor, the bandwidth into a light load is over 160mhz without peaking, but into a heavy load the bandwidth reduces to 100mhz. the loading has so much effect because there is a mild resonance in the output stage that enhances the bandwidth at light loads but has its q reduced by the heavy load. this enhancement is only useful at low gain settings; at a gain of ten it does not boost the bandwidth. at unity gain, the enhancement is so effective the value of the feedback resistor has very little effect. at very high closed-loop gains, the bandwidth is
9 lt1229/lt1230 amplifier at 150 c is less than 7ma and typically is only 4.5ma. the power in the ic due to the load is a function of the output voltage, the supply voltage and load resistance. the worst case occurs when the output voltage is at half supply, if it can go that far, or its maximum value if it cannot reach half supply. for example, lets calculate the worst case power dissipa- tion in a video cable driver operating on 12v supplies that delivers a maximum of 2v into 150 w . capacitive loads the lt1229/lt1230 can drive capacitive loads directly when the proper value of feedback resistor is used. the graph maximum capacitive load vs feedback resistor should be used to select the appropriate value. the value shown is for 5db peaking when driving a 1k load at a gain of 2. this is a worst case condition; the amplifier is more stable at higher gains and driving heavier loads. alterna- tively, a small resistor (10 w to 20 w ) can be put in series with the output to isolate the capacitive load from the amplifier output. this has the advantage that the amplifier bandwidth is only reduced when the capacitive load is present, and the disadvantage that the gain is a function of the load resistance. power supplies the lt1229/lt1230 amplifiers will operate from single or split supplies from 2v (4v total) to 15v (30v total). it is not necessary to use equal value split supplies, however, the offset voltage and inverting input bias current will change. the offset voltage changes about 350 m v per volt of supply mismatch, the inverting bias current changes about 2.5 m a per volt of supply mismatch. power dissipation the lt1229/lt1230 amplifiers combine high speed and large output current drive into very small packages. be- cause these amplifiers work over a very wide supply range, it is possible to exceed the maximum junction temperature under certain conditions. to ensure that the lt1229 and lt1230 remain within their absolute maximum ratings, we must calculate the worst case power dissipation, define the maximum ambient temperature, select the appropriate package and then calculate the maximum junction temperature. the worst case amplifier power dissipation is the total of the quiescent current times the total power supply voltage plus the power in the ic due to the load. the quiescent supply current of the lt1229/lt1230 has a strong nega- tive temperature coefficient. the supply current of each u s a o pp l ic at i wu u i for atio now if that is the dual lt1229, the total power in the package is twice that, or 0.602w. we now must calcu- late how much the die temperature will rise above the ambient. the total power dissipation times the thermal resistance of the package gives the amount of tempera- ture rise. for the above example, if we use the so8 surface mount package, the thermal resistance is 150 c/w junction to ambient in still air. temperature rise = p d (max) r q ja = 0.602w ? 150 c/w = 90.3 c the maximum junction temperature allowed in the plastic package is 150 c. therefore, the maximum ambient al- lowed is the maximum junction temperature less the temperature rise. maximum ambient = 150 c C 90.3 c = 59.7 c note that this is less than the maximum of 70 c that is specified in the absolute maximum data listing. if we must use this package at the maximum ambient we must lower the supply voltage or reduce the output swing. as a guideline to help in the selection of the lt1229/ lt1230 the following table describes the maximum sup- ply voltage that can be used with each part in cable driving applications. pvi vv v r pvmavv v w per amp d max s s max s o max o max l d max () () () () () =+ ? ? ? =+ () =+= 2 2 12 7 12 2 2 150 0 168 0 133 0 301 ... w
lt1229/lt1230 10 assumptions: 1. the maximum ambient is 70 c for the commercial parts (c suffix) and 125 c for the full temperature parts (m suffix). 2. the load is a double-terminated video cable, 150 w . 3. the maximum output voltage is 2v (peak or dc). 4. the thermal resistance of each package: j8 is 100 c/ w j is 80 /w n8 is 100 c/w n is 70 /w s8 is 150 c/w s is 110 /w maximum supply voltage for 75 w cable driving applications at maximum ambient temperature part package max power at t a max supply lt1229mj8 ceramic dip 0.500w at 125 cv s < 10.1 lt1229cj8 ceramic dip 1.050w at 70 cv s < 18.0 lt1229cn8 plastic dip 0.800w at 70 cv s < 15.6 lt1229cs8 plastic so8 0.533w at 70 cv s < 10.6 lt1230mj ceramic dip 0.625w at 125 cv s < 6.6 lt1230cj ceramic dip 1.313w at 70 cv s < 13.0 lt1230cn plastic dip 1.143w at 70 cv s < 11.4 lt1230cs plastic so14 0.727w at 70 cv s < 7.6 slew rate the slew rate of a current feedback amplifier is not independent of the amplifier gain the way it is in a tradi- tional op amp. this is because the input stage and the output stage both have slew rate limitations. the input stage of the lt1229/lt1230 amplifiers slew at about 100v/ m s before they become nonlinear. faster input sig- nals will turn on the normally reverse-biased emitters on the input transistors and enhance the slew rate signifi- cantly. this enhanced slew rate can be as much as 2500v/ m s. the output slew rate is set by the value of the feedback resistors and the internal capacitance. at a gain of ten with a 1k feedback resistor and 15v supplies, the output slew rate is typically 700v/ m s and C1000v/ m s. there is no input stage enhancement because of the high gain. large-signal response, a v = 2, r f = r g = 750 w u s a o pp l ic at i wu u i for atio lt1229 ? ta07 settling time the characteristic curves show that the lt1229/lt1230 amplifiers settle to within 10mv of final value in 40ns to 55ns for any output step up to 10v. the curve of settling to 1mv of final value shows that there is a slower thermal contribution up to 20 m s. the thermal settling component comes from the output and the input stage. the output contributes just under 1mv per volt of output change and the input contributes 300 m v per volt of input change. fortunately, the input thermal tends to cancel the output thermal. for this reason the noninverting gain of two configurations settles faster than the inverting gain of one. larger feedback resistors will reduce the slew rate as will lower supply voltages, similar to the way the bandwidth is reduced. large-signal response, a v = 10, r f = 1k, r g = 110 w lt1229 ? ta08
11 lt1229/lt1230 crosstalk and cascaded amplifiers the amplifiers in the lt1229/lt1230 do not share any common circuitry. the only thing the amplifiers share is the supplies. as a result, the crosstalk between amplifiers is very low. in a good breadboard or with a good pc board layout the crosstalk from the output of one amplifier to the input of another will be over 100db down, up to 100khz and 65db down at 10mhz. the following curve shows the crosstalk from the output of one amplifier to the input of another. amplifier crosstalk vs frequency frequency (hz) 10 50 output to input crosstalk (db) 60 70 80 90 100 120 100 1k 10k 100m lt1229 ?ta12 100k 1m 10m 110 v s = 15v a v = 10 r s = 50 w r l = 100 w u s a o pp l ic at i wu u i for atio the high frequency crosstalk between amplifiers is caused by magnetic coupling between the internal wire bonds that connect the ic chip to the package lead frame. the amount of crosstalk is inversely proportional to the load resistor the amplifier is driving, with no load (just the feedback resistor) the crosstalk improves 18db. the curve shows the crosstalk of the lt1229 amplifier b output (pin 7) to the input of amplifier a. the crosstalk from amplifier as output (pin 1) to amplifier b is about 10db better. the crosstalk between all of the lt1230 amplifiers is as shown. the lt1230 amplifiers that are separated by the supplies are a few db better. when cascading amplifiers the crosstalk will limit the amount of high frequency gain that is available because the crosstalk signal is out of phase with the input signal. this will often show up as unusual frequency response. for example: cascading the two amplifiers in the lt1229, each set up with 20db of gain and a C3db bandwidth of 65mhz into 100 w will result in 40db of gain, but the response will start to drop at about 10mhz and then flatten out from 20mhz to 30mhz at about 0.5db down. this is due to the crosstalk back to the input of the first amplifier. for best results when cascading amplifiers use the lt1229 and drive amplifier b and follow it with amplifier a. u s a o pp l ic at i ty p i ca l single 5v supply cable driver for composite video this circuit amplifies standard 1v peak composite video input (1.4v p-p ) by two and drives an ac coupled, doubly terminated cable. in order for the output to swing 2.8v p-p on a single 5v supply, it must be biased accu- rately. the average dc level of the composite input is a function of the luminance signal. this will cause problems if we ac couple the input signal into the amplifier because a rapid change in luminance will drive the output into the rails. to prevent this we must establish the dc level at the input and operate the amplifier with dc gain. the transistors base is biased by r1 and r2 at 2v. the emitter of the transistor clamps the noninverting input of the amplifier to 1.4v at the most negative part of the input (the sync pulses). r4, r5 and r6 set the amplifier up with a gain of two and bias the output so the bottom of the sync pulses are at 1.1v. the maximum input then drives the output to 3.9v. lt1229 ?ta11 + 1/2 lt1229 v out r3 150k r2 2k v in r5 750 w c1 1 m f r8 10k r1 3k c2 1 m f r4 1.5k 2n3904 5v c3 47 m f r6 510 w r7 75 w c4 1000 m f + +
lt1229/lt1230 12 u package d e sc r i pti o j8 package 8-lead cerdip (narrow .300 inch, hermetic) (reference ltc dwg # 05-08-1110) j package 14-lead cerdip (narrow .300 inch, hermetic) (reference ltc dwg # 05-08-1110) obsolete packages j8 1298 0.014 ?0.026 (0.360 ?0.660) 0.200 (5.080) max 0.015 ?0.060 (0.381 ?1.524) 0.125 3.175 min 0.100 (2.54) bsc 0.300 bsc (0.762 bsc) 0.008 ?0.018 (0.203 ?0.457) 0 ?15 0.005 (0.127) min 0.405 (10.287) max 0.220 ?0.310 (5.588 ?7.874) 12 3 4 87 65 0.025 (0.635) rad typ 0.045 ?0.068 (1.143 ?1.727) full lead option 0.023 ?0.045 (0.584 ?1.143) half lead option corner leads option (4 plcs) 0.045 ?0.065 (1.143 ?1.651) note: lead dimensions apply to solder dip/plate or tin plate leads j14 1298 0.045 ?0.065 (1.143 ?1.651) 0.100 (2.54) bsc 0.014 ?0.026 (0.360 ?0.660) 0.200 (5.080) max 0.015 ?0.060 (0.381 ?1.524) 0.125 (3.175) min 0.300 bsc (0.762 bsc) 0.008 ?0.018 (0.203 ?0.457) 0 ?15 1 234 56 7 0.220 ?0.310 (5.588 ?7.874) 0.785 (19.939) max 0.005 (0.127) min 14 11 8 9 10 13 12 0.025 (0.635) rad typ note: lead dimensions apply to solder dip/plate or tin plate leads
13 lt1229/lt1230 n8 package 8-lead pdip (narrow .300 inch) (reference ltc dwg # 05-08-1510) s8 package 8-lead plastic small outline (narrow .150 inch) (reference ltc dwg # 05-08-1610) u package d e sc r i pti o n8 1098 0.100 (2.54) bsc 0.065 (1.651) typ 0.045 ?0.065 (1.143 ?1.651) 0.130 0.005 (3.302 0.127) 0.020 (0.508) min 0.018 0.003 (0.457 0.076) 0.125 (3.175) min 12 3 4 87 6 5 0.255 0.015* (6.477 0.381) 0.400* (10.160) max 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.035 0.015 +0.889 0.381 8.255 () *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.010 inch (0.254mm) 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) so8 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 8 7 6 5 0.189 ?0.197* (4.801 ?5.004) 0.228 ?0.244 (5.791 ?6.197) dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
lt1229/lt1230 14 s package 14-lead plastic small outline (narrow .150 inch) (reference ltc dwg # 05-08-1610) u package d e sc r i pti o 1 2 3 4 0.150 ?0.157** (3.810 ?3.988) 14 13 0.337 ?0.344* (8.560 ?8.738) 0.228 ?0.244 (5.791 ?6.197) 12 11 10 9 5 6 7 8 0.016 ?0.050 (0.406 ?1.270) 0.010 ?0.020 (0.254 ?0.508) 45 0 ?8 typ 0.008 ?0.010 (0.203 ?0.254) s14 1298 0.053 ?0.069 (1.346 ?1.752) 0.014 ?0.019 (0.355 ?0.483) typ 0.004 ?0.010 (0.101 ?0.254) 0.050 (1.270) bsc dimension does not include mold flash. mold flash shall not exceed 0.006" (0.152mm) per side dimension does not include interlead flash. interlead flash shall not exceed 0.010" (0.254mm) per side * **
15 lt1229/lt1230 information furnished by linear technology corporation is believed to be accurate and reliable. however, no responsibility is assumed for its use. linear technology corporation makes no represen- tation that the interconnection of its circuits as described herein will not infringe on existing patent rights. n package 14-lead pdip (narrow .300 inch) (reference ltc dwg # 05-08-1510) u package d e sc r i pti o n14 1098 0.020 (0.508) min 0.125 (3.175) min 0.130 0.005 (3.302 0.127) 0.045 ?0.065 (1.143 ?1.651) 0.065 (1.651) typ 0.018 0.003 (0.457 0.076) 0.100 (2.54) bsc 0.005 (0.125) min 0.255 0.015* (6.477 0.381) 0.770* (19.558) max 3 1 2 4 5 6 7 8 9 10 11 12 13 14 0.009 ?0.015 (0.229 ?0.381) 0.300 ?0.325 (7.620 ?8.255) 0.325 +0.035 0.015 +0.889 0.381 8.255 () *these dimensions do not include mold flash or protrusions. mold flash or protrusions shall not exceed 0.010 inch (0.254mm)
lt1229/lt1230 16 ? linear technology corporation 1992 linear technology corporation 1630 mccarthy blvd., milpitas, ca 95035-7417 (408) 432-1900 l fax: (408) 434-0507 l www.linear.com 122930fb lt/cp 0801 1.5k rev b ? printed in usa related parts part number description comments lt1227 single 140mhz cfa single version of the lt1229 lt1395/lt1396/lt1397 single/dual/quad 400mhz cfa sot-23, msop-8 and ssop-16 packaging u s a o pp l ic at i ty p i ca l single supply ac coupled amplifiers noninverting inverting lt1229 ?ta10 + + 1/2 lt1229 5v 4.7 m f 0.1 m f 510 w v out 10k w 10k w v in a v = 10 bw = 600hz to 50mhz 51 w r s + 510 w r s + 51 w 4.7 m f lt1229 ?ta09 + + 1/2 lt1229 5v 4.7 m f 0.1 m f 510 w 51 w v out 10k 10k v in a v = 11 bw = 600hz to 50mhz + 4.7 m f

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