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APPLICATION NOTE
TEA5101A - RGB HIGH VOLTAGE AMPLIFIER BASIC OPERATION AND APPLICATIONS
By Ch. MATHELET
SUMMARY
I I.1 I.2 I.3 I.4 I.4.1. I.4.2. I.4.3. II II.1 II.1.1 II.1.2. II.1.2.1. II.1.2.2. II.2 II.2.1. II.2.2. III III.1 III.1.1. III.1.2. III.1.3. III.1.4. III.2 III.2.1. III.2.2. III.2.2.1. III.2.2.2. III.2.2.3. III.2.3. III.2.3.1. III.2.3.2. IV IV.1 IV.2 IV.3 IV.4 IV.5 V V.1 V.2 V.2.1. V.2.2. V.2.2.1. V.2.2.2. V.2.2.3. DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . INPUT STAGE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . OUTPUT STAGE . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . BEAM CURRENT MONITORING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . PROTECTION CIRCUITS. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . MOS Protection. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Protection Against Electrostatic Discharges . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Flashover Protection . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . FUNCTIONAL DESCRIPTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . VOLTAGE AMPLIFIER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Bias Conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Dynamic Operation . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . White To Black Transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Black To White Transition . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . BEAM CURRENT MONITORING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Stationary State. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Transient Phase . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . EXTERNAL COMPONENTS CALCULATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . COMPONENTS VALUE CALCULATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Feedback resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Input resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Bias resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Current measurement resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . DISSIPATED POWER . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement method . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Static power. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement with sinusoidal input . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurement in a TV set . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Design of external components. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Heatsink . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Power rating of feedback resistor . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . APPLICATION HINTS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . DYNAMIC PERFORMANCES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . CROSSTALK . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . FLASHOVER PROTECTION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . OUTPUT SWING . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . LOW CURRENT MEASUREMENTS . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . APPLICATION EXAMPLES . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . APPLICATION DESCRIPTION. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . PERFORMANCES EVALUATION . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Measurements conditions . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Results . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Bandwidth . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Crosstalk . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . Transition times. . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . .
Page
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AN377/0594
TEA5101A APPLICATION NOTE
The aim of this Application Note is to describe the basic operation of the TEA5101A video amplifier and to provide the user with basic hints for the best utilization of the device and the realisation of high performance applications. Application examples are also provided to assist the designer in the maximum exploitation of the circuit. GENERAL The control of state-of-the-art color cathode ray tubes requires high performance video amplifiers which must satisfy both tube and video processor characteristics. When considering tube characteristics (see Figures 13 and 14),we note that a 130V cutoff voltage is necessary to ensure a 5mA peak current.However 150V is a more appropriate value if the saturation effect of the amplifier is to be taken into account. As the dispersion range of the three guns is 12%, the cutoff voltage should be adjustable from 130V to 170V. The G2 voltage, from 700 to 1500V allows overall adjustment of the cutoff voltage for similar tube types. A 200V supply voltage of the video amplifier is necessary to achieve a correct blanking operation. In addition, the video amplifier should have an output saturation voltage drop lower than 15V, as a drive voltage of 130V (resp. 115V) is necessary to obtain a beam current of 4 mA for a gun which has a cutoff point of 170V (resp. 130V). Note : For all the calculations discussed above, the G1 voltage is assumed to be 0V. The video processor characteristics must also be considered. As it generally delivers an output voltage of 2 to 3V, the video amplifier must provide a closed loop DC gain of approximately 40. The video amplifier dynamic performances must also meet the requirements of good definition even with RGB input signals (teletext,home computer...), e.g. 1mm resolution on a 54cm CRT width scanned in 52s. Consequently, a slew rate better than 2000V/s, i.e. rise and fall times lower than 50ns, is needed. In addition, transition times must be the same for the three channelsso as to avoid coloured transitions when displaying white characters. The bandwidth of a video amplifier satisfying all these requirements must be at least 7MHz for high level signals and 10MHz for small signals. One major feature of a video amplifier is its capability to monitor the beam current of the tube. This function is necessary with modern video processors: - for automaticadjustmentof cutoff and also, where required,video gain in order to improve the long term performances by compensation for aging effects through the life of the CRT. This adjustment can be done either sequentially (gun after gun) or in a parallel mode. - for limiting the average beam current A video amplifier must also be flashover protected and provide high crosstalk performances. Crosstalk effects are mainly caused by parasitic capacitors and thus increase with the signal frequency. A crosstalk level of -20dB at 5MHz is generally acceptable. Table 1 summarizes the main features of a high performance video amplifier. Table 1 : Main Features of a High Performance Video Amplifier
Maximum Supply Voltage Output voltage swing "Average" Output voltage swing "Peak" Low level saturation (refered to VG1) Closed loop gain Transition time Large signal bandwidth Small signal bandwidth Beam current monitoring Flash over protection Crosstalk at 5MHz -20dB 220V 100V 130V 15V 40 50ns 7MHz 10MHz
The SGS-THOMSON Microelectronics TEA5101A is a high performance and large bandwidth 3 channel video amplifier which fulfills all the criteria discussed above. Designed in a 250V DMOS bipolar technology, it operates with a 200V power supply and can deliver 100V peak-to-peak output signals with rise and fall times equal to 50ns. The 5101A features a large signal bandwidth of 8MHz, which can be extended to 10MHz for small signals (50 Vpp). Each channel incorporates a PMOS transistor to monitor the beam current. The circuit provides internal protection against electrostatic discharges and high voltage CRT discharges. The best utilization of the TEA5101Ahigh performance features such as dynamic characteristics, crosstalk,or flashover protection requires optimized application implementation. This aspect will be discussed in the fourth part of this document.
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TEA5101A APPLICATION NOTE
I - DESCRIPTION The complete schematic diagram of one channel of the TEA5101A is shown in Figure 1. Figure 1
V DD 5 40k 13 20k (10, 7) 15 (12, 9)
0.8k 14 (11, 6) 1k
2 2.5k 1 (3, 4) 35 35 6k 3pF
350 1.5k
5101A-02.EPS
GND 8
I.1 - Input Stage The differentialinput stage consists of the transistor T1 and T2 and the resistors R4,R5 and R6. This stage is biased by a voltage source T3,R1,R2 and R3. R3 Each amplifier is biased by a separate voltage source in order to reduce internal crosstalk. The load of the input stage is composed of the transistor T4 (cascode configuration) and the resistor R7. The cascode configuration has been chosen so as to reduce the Miller input capacitance. The voltage gain of the input stage is fixed by R7 and the emitter degenerationresistors R5,R6,and the T1,T2 internal emitter resistances. The voltage gain is approximately 50dB. Using a bipolar transistor T4 and a polysilicon resistor R7 gives rise to a very low parasitic capaciVB(T1) = (1 +
tance at the output of this stage (about 1.5pF). Hence the rise and fall times are about 50ns for a 100V peak-to-peak signal (between 50V and 150V). I.2 - Output Stage The output stage is a quasi-complementary class B push-pullstage. This design ensuresa symetrical load of the first stage for both rising and falling signals. The positive output stage is made of the DMOS transistor T5,and the negative output stage is made of the transistors PMOS T6 and DMOS T7. The compound configuration T6-T7 is equivalent to a single PMOS. A single PMOS transistor capable of sinking the total current would have been too large. By virtue of the symetrical drive properties of the output stage the rise and fall times are equal (50ns for 100V DC output voltage).
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R2
) x VB(T3) 3.8V
TEA5101A APPLICATION NOTE
I.3 - Beam Current Monitoring This function is performed by the PMOS transistor T8 in source follower configuration. The voltage on the source (cathode output) follows the gate voltage (feedback output). The beam current is absorbed via T8 . On the drain of T8, this current will be monitored by the videoprocessor. I.4 - Protection Circuits I.4.1 - MOS protection Four zener diodes DZ(1-4) are connected between gate and source of each MOS in order to prevent the voltage from reaching the breakdown voltage.Hence the VGS voltage is internally limited to 15V. I.4.2 - Protection against electrostatic discharges All the input/outputpins of the TEA5101A are protected by the diodes D1-D7 which limit the overvoltage due to ESD. I.4.3 - Flashover Protection A high voltage and high current diode D5 is connected between each output and the high voltage power supply. During a flash, most of the current is generally absorbed by the spark gap connected to the CRT socket. The remaining current is absorbed by the high voltage decoupling capacitor through the diode D5. Hence the cathode voltage is clamped to the supply voltage and the output voltage does not exceed this value.
II - FUNCTIONAL DESCRIPTION The schematic diagram of one TEA5101A channel with its associated external components is shown in Figure 2. Figure 2
Rf VDD 5 R7 40k T5 Dz1 Dz2 R9 0.8k T6 T7 D4 R8 1k VCC
2
9
Feedback Output (12, 15)
C
D5 D3 Dz3 T8 D6 D7
6
Dz4
7
R10 20k
(10, 13)
CL CRT
Rm
t0 Video Processor
T4 R1 2.5k T1 R5 35 R6 35 R4 350 T3 R3 1.5k T2 R2 6k C 3pF
D1 Re Input
1
(3, 4)
Rp
D2
8
GND
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5101A-04.EPS
TEA5101A APPLICATION NOTE
II.1 - Voltage Amplifier II.1.1 - Bias conditions Vin = Vref The bias point is fixed by the feedback resistor Rf,the bias resistor Rp, and by the internal reference voltage when Vin = Vref. If VO is the output voltage (pin 9) : Rf VO = (1 + ) x Vref (1) Rp In this state T1 and T 2 are conducting. A current flows in R7 and T4 soT5 is on. The T5 drain current is fed to the amplifier input through the feedback resistor. The current in R7 is: VDD - VO - VGS(T 5) VDD - VO I(R7) = R7 R7 and the current in T5 and Rf is : VO - Vref VO I(T5) = Rf Rf Thus the total current absorbed by each channel of the TEA5101A is : VDD 1 1 + VO x ( - ) R7 Rf R7 The cathode (pin 7) output voltage is: VO + VGS(T8) = VO The beam current is absorbed by T8 and Rm. The voltage developed across Rm by this current is fed to the videoprocessor in order to monitor the beam current. II.1.2 - Dynamic operation The TEA5101Aoperates as a closed loop amplifier, with its voltage gain fixed by the resistors Rf and Re. Since the open loop gain A isnot infinite, the resistor Rp and the input impedance Rin must be considered.Hence the voltage gain is Rf 1 (2) x G=- Rf Re 1 ) 1 + (1 + Rp Re Rin A II.1.2.1 - Input voltage Vin < Vref (black picture) In this case the current flowing in R7 and T1 decreases whilst the collector voltage of T4 and the output voltage both increase. In the extreme case, I(T1) = I(R7) = 0 and VO= VDD-VGS(T5) In order to charge the tube capacitor the voltage is fed to the cathode output in two ways: - through the PMOS (with a VGS difference) for the low frequency part - through the capacitor C for the high frequency part (output signal leading edge) To correctly transmit the rising edge, the value of the capacitor C must be high compared to C L. With the current values used (C = 1nF,CL = 10pF), the attenuation is very small (0.99) II.1.2.2 - Input voltage Vin > Vref (white picture) In this case,the current in R7 and T1 increases with an accompanyingdrop of T4's collector voltage until T1 and T4 are saturated. At this point: VO VC(T4) VCC During a high to low transition (i.e. black-white picture), the beam current is absorbed in two ways: - through the capacitor C and the compound PMOS T6-T7 for the high frequency part (falling edge) - through the PMOS T8 and the resistor Rm for the low frequency part. II.2 - Beam Current Monitoring II.2.1 - Stationary state The beam current monitoring is performed by the PMOS T8 and the resistor Rm. When measuring low currents (leakage, quasi cutoff),the Rm value is generally high. When measuring high currents (drive, average or peak beam current),Rm is generally bypassed by a lower impedance. It should be noted that the current supplied by the three guns flows through this resistor.Hence,with too large a value for the resistor Rm,the cathode voltage of the tubes will become too high for the required operating current values.This is a fundamental difference between the TEA5101A and discrete video amps. In discrete video amps, the current monitoring transistor is a high voltage PNP bipolar which may saturate. In this case the beam current can flow through the transistor base and it is no longer monitored by the video processor. This effect does not occur with the TEA 5101A. II.2.2 - Transient phase : low current measurements The cut-off adjustment sequence is generally as follows: In a first step, the cathode is set to a high voltage (180V) in order to blank the CRT and to measure the leakage current. In a second step, the tube is slighly switched on to measure a very low current (quasi cut-off current). This operation is performed by setting the cathode voltage to about 150V and adjusting it until the proper current is obtained. The maximum time available to do this operation is generally about 52s. Figure 3 shows the simplified diagram of the TEA5101A output, the voltages during the different steps,and the stationary state the system must reach for correct adjustment.
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TEA5101A APPLICATION NOTE
Figure 3
2 VC 1.5V 180V 2.5V
9
150V C 1nF VC 181.5V
7
152.5V 151.5V 181.5V K CL 1 = R x C = 10ns 2 = R x C = 1s 152.5V 151.5V CUT-OFF
5101A-05.EPS
R 1k
BLANKING
During the blanking phase, the tube is switched off, the PMOS is switched off and its VGS voltage is equal to the pinch-off voltage (about 1.5V). The voltages at the different nodes are shown in figure 3 (V(9) = 180V, V(k) = 181.5V). The falling edge of the cutoff pulse is instantaneously transmitted by the capacitor C. When the stationary state is reached, the cathode voltage will be 152.5V if the voltage on pin 9 is 150V, as the VGS voltage of the conducting PMOS is about 2.5V. We can see that the voltage on C must increase by an amount of Vc = 1V. This charge is furnished by the tube capacitor which is discharged by an amount of VCL = 29V with a time constant equal to R x CL (10 ns). By considering the energy balance, we can calculate the maximum charge Vmax that CL can furnished to C
Vmax = C L x VCL C 3V
reached without any contribution being required from the tube current,i.e. the whole tube current can flow throughthe PMOS and the adjustmentcan be performed correctly. Considering higher voltage and beam current swings, the margin is greater because: - the voltage swing across the tube capacitor is greater - the tube current is higher and the picture is not disturbed even if part of the beam current is used to charge the capacitor C. III - EXTERNAL COMPONENTS CALCULATION The implementation of the TEA5101A in an application requires the determinationof external component values. These components are Rf, Re, Rp and Rm (see Figure 4). The dissipated power in the IC and in the feedback resistor Rf must also be calculated in order to correctly choose the power ratings of the heatsink and resistors.
Since this voltage is greater than VC, the capacitor C can be charged and the stationary state is
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TEA5101A APPLICATION NOTE
Figure 4
VIN(BLK) Re
4
Rf
9
VOUT(BLK) C
Rp
VREF
6
7
VIDEO PROC. VC
I C.O.
5101A-06.EPS
Rm
III.1 - Components Value Calculation From equations 1 and 2 in section II-1, both the value of the DC output voltage and the voltage gain depend directly on the resistor Rf. Hence Rf must be determined first before calculating the value of Re and Rp in order to obtain the correct gain and DC output voltage. III.1.1 - Feedback resistor Rf The value of Rf must be as low as possible in order to obtain the optimum dynamic performance from the TEA5101A(see section IV-1). A typical value of Rf is 39 k. III.1.2 - Input resistor Re The voltage gain is calculated from the following formula (see section II-1):
G=- Rf
level output voltage VOUT(BLK) is equal to the cutoff voltage, which is a characteristic of the tube currently used, when the DC black level input voltage VIN(BLK) is the mean value of the adjustment range of the video processor. This is the optimum condition to ensure a correct adjustment during the lifetime of the tube. Rp can be calculated by considering the TEA5101Aas an operational amplifier and applying the usual formula :
Rp = Vref Vout (BLK) - Vref Vin (BLK) - Vref + Rf Re
- If Vin(BLK) = Vref
Rp =
Vref x Rf Vout (BLK) - Vref
Re
1 Rf 1 1+ (1 + ) A R p Re R in
Since the open loop gain A is high enough (50dB), we can approximate the calculation:
G@- Rf
Re
where Re is generally implemented as a variable value for channel gain adjustment. If the gain adjustment range Gmin, Gmax is known:
R e min = Rf
For a 150V black level : Rp = 1k with Rf = 39k - If Vin (BLK) Vref : Vin (BLK) = 2.7V Rp = 1.2k with Rf = 39k Re = 1.5k Or Vin (BLK) = 6.7V with Rf = 39k Rp = 680 Re = 1.5k for a 150V black level III.1.4 - Current measurement resistor Rm Rm must be determined by taking into account the quasi cutoff current Ico and the input voltage VC of the video processor. VC Rm = ICO - With the videoprocessor TEA5031D (VC = 2V) : Rm = 120k with ICO = 16A
Gmax
and Re max =
Rf
Gmin
With Gmin = 15 and Gmax = 80 : Re will be made of a 2.2k potentiometerand 470 fixed resistor. III.1.3 - Bias resistor Rp Rp must be chosen in such a way that the black
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TEA5101A APPLICATION NOTE
- With the videoprocessor TDA3562A (VC = 0.5V) which requires a DC biased input "Black current stabilization" (pin 18), the schematic diagram is the following : Figure 5
7
12V
6
120k TDA3562A Pin 18 82k I C.O
The DC bias is 12 x
0.5 (
82 = 5V 120 + 82 The quasi cutoff current is
1
120 +
1
82
) x 1 x 10-3 = 10A
When the IC is driven by a sinusoidal signal (capacitive drive),the measurement and calculation are straightforward : - VOUT(AVG) = VOUT(DC) VOUT (peak to peak) - VOUT(RMS) = 2 x 2 With VOUT (DC) = 100V and VOUT (peak to peak) = 100V and Rf = 39k PSR = 0.8W PDR = 0.1W Measurements are more difficult to carry out when the IC is working in a TV set. VOUT(AVG) can be measured with an oscilloscope (difference of level between AC and DC coupling) andVOUT (RMS) can be measured by connecting an RMS voltmeter to the feedback resistor. In this case we have the following results (see section 2.2.3) : - VOUT (AVG) = 130V and PSR = 1.3W - VOUT (RMS) = 32V and PDR = 80mW In each case, the term PDR can be neglected as a reasonable approximation.Hence, the power dissipated by the IC will be: 3V2 OUT (AVG) Rf and the power dissipated in Rf will be : Pi = VDD x IDD Pr =
III.2 - Dissipated Power in External Components The only components dissipating power are the TEA5101A and the feedback resistor. The dissipated power has a constant static component and a dynamic component which increases with frequency. The theoretical calculation is not sufficiently accurate to determine the correct dissipated power. The best way consists of measuring the power in different configurations of the circuit: steady state (no input), sinusoidal input,and in situ (in a TV set with a video input signal). The measurement method will be described first and then the results and calculations will be discussed. III.2.1 - Measurement method The dissipated power can be determined by measuring the average supply current IDD (principally high voltage supply current VDD) and by subtracting the power dissipated in the external components from the calculated power delivered by this supply voltage. The power delivered by the high voltage power supply is : P = VDD x IDD The power dissipated in the external components (principally the feedback resistor Rf) is : 3 x V2 OUT (AVG) - for the static part: PSR = Rf 3 x V2 OUT (RMS) - for the dynamic part: PDR = Rf
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5101A-07.EPS
V2 OUT (AVG) Rf
III.2.2 - Results III.2.2.1 - Static power Table 2 shows the measured values of IDD and the calculated power for three values of Vout and for VDD = 200V Table 2
VOUT (V) 50 100 150 IDD (mA) 16 15 14.6 Pi (W) 3 2.2 1.2 Pr (W) 0.065 0.25 0.6
We can see that the static power dissipated in the IC decreases with VOUT increasing, but obviously the power dissipated by Rf increases as VOUT increases. III.2.2.2 - Measurement with sinusoidal input Table 3 summarizes the results obtained from practical measurements as functions of VOUT(DC) and of the frequency (the three channels are driven simultaneously). We can see that when driving the IC with a HF sinusoidal signal, care must be taken to avoid excessive temperature increase.
TEA5101A APPLICATION NOTE
Table 3
VOUT (V) 50 100 150 IDD 1MHz (mA) 20.7 20 18 IDD 7MHz (mA) 44.6 59.5 45 VOUT (PP) 1MHz (V) 66 100 100 VOUT (PP) 7MHz (V) 50 80 67 Pi 1MHz (W) 3.9 3 1.7 Pi 7MHz (W) 8.7 11 8.2 Pr (W) 0.065 0.25 0.6
VOUT VDD I (AVG) DD (mA) (V) (V) Bright.max Noise Bright.min Bright.max Multiburst Bright.min 148 188 131 158 22.2 23.3 23.6 22 218 224 213 221
Pi (W)
Pr (W)
3.15 0.56 2.5 0.9 3.7 2.9 0.44 0.64
Cf is the parasitic capacitor between the input and the output. Cin is the parasitic capacitor between the input and ground. The voltage gain versus frequency can be deduced from the formula (2) in chapter II section 1.2 :
G(s) = Rf R e (1 + Rf Cf s) 1 Rf 1 +R eq Cin s 1 1+ 1+ A(s) R eq 1 + R f Cf s
III.2.3 - Design of heatsink and external components III.2.3.1 - Heatsink As discussed above, the power dissipated in the IC in a TV set can reach about 4W. In this case, a 12oC/W heatsink seems to be sufficient. Such a heatsink will give Tj = 115oC for Troom = 60oC. The resulting margin guarantees correct reliability. III.2.3.2 - Feedback resistors 1 Watt type feedback resistors must be used, as they may need to dissipate 0.9W when the TV set is working and up to 1W when the TV is blanked (VOUT = 200V), for example when the security of the scanning processor is activated. IV - APPLICATION HINTS IV.1 - Dynamic Performances Figure 6 shows the simplified schematic diagram of the TEA5101A in AC mode.
with Req=Rp//Re//Rin and A(s) open loop gain A(s) is a se con d order f un ctio n su ch a s AO 1 + bs + as2 with a = 9 x 10-16 s2, b = 60 x 10-9 s , AO = 400 Assuming Req x Cin = Rf x Cf, we find:
G(s) = -
Rf
Re (1 + Rf Cf s)
x
1 x B B B 1+ as2 1+ bs + AO AO + B AO + B
1
Rf Req We see that the closed loop amplifier is equivalent to a combination of a second order circuit and a first order one. The latter comprises the feedback resistor and the parasitic capacitor between input and output. With the current values : Rf = 39k, Re = 2k, Rin = 14k, Rp = 1.2k, Cf = 0.5pF, Cin = 15pF with B = 1 +
9/21
5101A-08.EPS
III.2.2.3 - Measurement in a TV set We have determined the worst cases of dissipation in a TV set. These trials have been carried out on one particular TV set, and may not be representative for all TV sets. In this particular TV set, the worst cases of dissipation occur with noise signal (from HF tuner) and with a multiburst pattern (0.8 to 4.8MHz) in RGB mode. Table 4 summarizes the results in these two cases when the brightness control is set to min and max value (the contrast control is set to max). Table 4
Figure 6
Cf Rf Re A(s) Rp C IN
TEA5101A APPLICATION NOTE
we have Req x Cin = 10ns, Rf x Cf = 20ns, B = 56 The second order circuit characteristics are : Natural frequency : 1 AO+ B Fn = x = 15MHz 2 xxa B damping factor : b B z= x = 0.35 2 x a AO + B The cut off frequency of the first order circuit is : 1 = 8MHz fC = 2 x Rf x Cf The amplifier response is thus the combination of the responses of these two circuits. The contribution of the parasitic capacitor Cf to the frequency response is very important. If the value of Cf is too high, the contribution of the first order circuit will be of overriding importance and the resulting bandwidth of the amplifier will be too small. If the value of Cf is too low, the response curve will have a peak (due to the second order circuit). A "ringing" effect will be present on pulse-type signals and an instability and oscillation can occur at somefrequencies. This capacitor is generally too high. It consists of: - the self parasitic capacitor of the feedback resistor - the parasitic capacitor due to the PCB layout. Practically,the best bandwidth performances are achieved by: - the smallest input-output capacitor and the smallest capacitor between an input and ground - using a feedback resistor with the smallest possible value but large enough to yield a sufficiently high gain. - using a feedback resistor with small parasitic capacitance (typ 0.2pF). Some resistors have 0.5 or 0.8 pF parasitic capacitor. The parasitic capacitors discussed above are usually the ones which need to be taken into account. However any other parasitic capacitor or inductor can modify the frequency response. For instance,a too large capacitor value between the feedback output and ground can create a dominant pole and cause a potential risk of oscillation . IV.2 - Crosstalk Figure 7 shows the different parasitic links inducing crosstalk. The crosstalk can be caused by: - parasitic coupling between the inputs (Cpi) - parasitic coupling between the outputs (Cpo) - parasitic coupling between an output and a near input of another channel (Cp). Figure 7
Rf
i1 Cp C pi Rf
O1
C po
i2
O2
Parasitic coupling may be capacitive or be caused by HF radiations. The third type of parasitic coupling is predominant since it involves the addition by feedback at relatively high level(output) signals to relatively low level (input) signals. For example, a 0.1pF Cp parasitic capacitor between an output and the input of another channel will act as a differenciator with the feedback resistor Rf = 39k. The transfer function of this integrator will be Rf x Cp x s ( 0.2j at 8MHz) and thus the crosstalk will be -14dB at 8MHz. The parasitic coupling between inputs and outputs must be minimized to achieve an acceptable crosstalk (-20dB at 5MHz). This can be done by crossing only the input wires and separating the input and output leads. High voltage components and wires must be laid out as far as possible from small signal wires,even if this results in a larger circuit board. HF radiations from the feedback resistor must not induced a voltage signal at the input of another channel. This can be achieved by : - spacing out the feed back resistors - mounting these resistors in the same direction and strictly aligned one under another. - mounting these resistors 1cm above the PC board - using ground connections to insulate the input wires IV.3 - Flashover Protection A picture tube has generally several high voltage discharges in its lifetime. This is due to the fact that the vacuum is not perfect coupled with the presence of metallic particles evaporated from the electrodes.Hence, short circuits (very brief fortunately) can occur between two electrodes,one of which is usually the anode (at EHT potential). An overvoltage can be induced on the cathodes or on the
10/21
5101A-09.EPS
TEA5101A APPLICATION NOTE
supplies even if a flash occurs on an electrode other than a cathode,because of the possibility of flashes in series or overvoltages due to inductive links on the video board or on the chassis. These overvoltages can destroy an IC particularly the video amplifier which is the most vulnerable since it is directly connected to the tube. The tube manufacturers have made much progress in technology in order to reduce the frequency of flashes and their associated energy (increased quality of vacuum, internal resistance for "soft flash" tubes). Nevertheless, some protection measures are suggested by the tube manufacturers : - connect spark gaps on each electrode (1 to 3kV or 12kV for focus) - connect the spark gaps to a separated ground directly connected to the chassis ground by a non inductive link - connect the cathodes or grids by protective resistors.These resistors must be able to withstand 12kV (20kV for focus)instantaneous voltages without breakdown and without any change of value following successive flashes. These resistors must be of a non-capacitive type. 1/2W (1W for focus) hot molded carbon type resistors are well suited for this application. - the grid and cathode connections on the PC board must be as short as possible and spaced well away from other connections in order to avoid parasitic inductions. Furthermore, the TEA5101A has been provided with an additional effective feature to improve the flashover protection.As described in section I-4, a protection device has been included comprising a high voltage high current diode which is connected between each output and the high voltage power supply. The equivalent diagram of this protection is shown in Figure 8. Figure 8 The flash current is diverted to the ground through the diode and the decoupling capacitor C. Two kinds of flashes can occur: 1) low resistance flashes during which the spark gaps are activated since the cathode voltage exceeds the breakdown value of the spark gap.In this case the equivalent diagram is the following : Figure 9
If Lf R I D5
V
Ctube 1nF
Ve SPARK GAP 2KV
Cf
VC
If flash current ( 1000A) Lf inductance of the connection ( 10H)
Ctube previously charged to 28kV is instantaneously discharged during V t = Ctube x = 30ns If Since the voltage across the spark gap falls almost instantaneously to 2000V, the peak current I flowing into the diode is (assuming VC is held by good decoupling) : Ve x t = 6A I= Lf To ensure a variation of VC less than 10V, C must be Ixt eg C > 18nF C> VC The decoupling must have good HF characteristics. 2) high resistance flashes in which the spark gaps are not activated. In this case the equivalent diagram is the following : Figure 10
Rt
R 1k
7
K SPARK GAP 2KV
D5 TEA5101A
5
I
D5
VC
Ctube 1nF V
C
C
5101A-10.EPS
CHASSIS
11/21
5101A-12.EPS
VDD
5101A-11.EPS
TEA5101A APPLICATION NOTE
2000 , I < 2A and Rt 12k R The time constant of the flash is Rt x Ctube = 12 s, the decaytime is approximatevely 30 s. The value of C must be txI eg C > 6F C> VC in order to ensure a VC variation less than 10V. The total decoupling will be made up by a 10F electrolytic capacitor connected in parallel with a 22nF plastic film capacitor with good HF properties. It must be placed very close to the TEA5101Ato be efficient. Otherwise, the equivalent diagram will be the following (case of low resistance flash). If V < 2 kV, I < Figure 11
R D5
characteristic curves (Figures 13 and 14). The minimum value of Vk (due to all the voltage drops in the resistors and in the amplifier) is given by the equation (see Figure 12) : Vk = (R + Ron + R1 + 3 x Rm) x It = Req x It (1) with Ron : on state PMOS resistance Figure 12
C
R
K
It
It It TEA5101A R1 Rm CRT
5101A-14.EPS
I C
Ctube
SPARK GAP
L P1 1H
V DD 220V
L P2
L P2 CHASSIS
5101A-13.EPS
To find the maximum available current Itmax,we can draw the curves of the equation (1) on the tube characteristics. Itmax will be given by the intersection point of the curves.Since the tube characteristics are: It vs Vcutoff + VG1 - Vk the equation (1) must be changed to
It = VCUTOFF + VG1 - Vk R eq (2)
VC = 210V with Lp1 = 1 H and Lp2 = 0 In this case the VDD voltage can rise to a dangerous value (+210V increase) and the protection is not efficient. If the connection between the socket ground and the chassis ground is inductive (Lp2 0), the effect is the same. However in this case, all the TV IC's,and not only the TEA5101A,will be exposed to destructive overvoltages. IV.4 - Output Swing The simplified diagram of this function is shown in Figure 12 (see Chapter II and chapter III ). The current delivered by a CRT is given by the
I x t LP1 x I + VC = C t
Assuming VG1= 0, we can draw the curves of equation (2) for several values of Vcutoff (eg 150V a n d 2 0 0 V) a n d s e v era l va lu e s o f Req (eg 5k,10k,15k,20k) (see Figures 13 and 14). We can see from these curves that Req must have the following values to allow the tube to source 4mA per gun : for a 150V cutoff point Req 5k or Req 15k for a 200V cutoff point As Ron value is approximatively 1.7k, the measurement resistor must be as low as possible. Working with higher cutoff point would be an alternative solution. But a 200V cutoff point seems to be too high a value since in this case the supply voltage would be greater than 200V and would affect reliability performances.
12/21
TEA5101A APPLICATION NOTE
Figure 13
10 5
high current operation of the TEA5101A is similar to that of a discrete amplifier (with PNP) operation. Figure 15
ANODE CURRENT PER GUN (A)
Heater Voltage - 6.3V Anode-to-grid No 1 Voltage - 25kV Grid No 3 -to-grid No 1 Voltage (each gun) adjusted to provide spot cut-off -Zero - Bias point
10 4
15k
15 0V 200 V
10 5k k
VZ
7 9
It
20 k
6
100 V
10 3
EK
c1
VIDEO SIGNAL VOLTAGE PER GUN V = VCUT-OFF - VG1K (VCUT-OFF = 150V)
Figure 14
10 5
ANODE CURRENT PER GUN ( A)
Heater Voltage - 6.3V Anode-to-grid No 1 Voltage - 25kV Grid No 3 -to-grid No 1 Voltage (each gun) adjusted to provide spot cut-off -Zero - Bias point
10 4
20 k
100 V
VIDEO SIGNAL VOLTAGE PER GUN V = VCUT-OFF - VG1K (V CUT-OFF = 200V)
5101A-16.EPS
10 2 10
EK
cL = 5
10 3
0V
150V
10
2
200 V
15k 10k 5k
10
3
5101A-15.EPS
10 2 10
10
2
10
3
For low currents, if the zener voltage is greater than the VGS voltage, the zener diode is biased off and the beam current flows through the measurement resistor. When the cathode voltage (pin 7) drop is limited because of the pin 6 voltage and when the pin 9 voltage continues to decrease,the zener diode is switched on when V7 - V9 = VZ. In this case the beam current is absorbed by the voltage amplifier and the tube can provide larger current values.Nevertheless, the pin 7 output voltage will follow the pin 9 voltage with a VZ difference. Since the pin 9 voltage is internally limited to 14V, the output voltage will be limited to 22V with a 8V zener diode. The CRT bias voltages shown on the previous curves are referenced to the G1 voltage. The TEA5101Ais referenced to ground. We can choose to work with a G1 voltage greater than ground and thus the low level saturation is not taken into account. In this case, the cutoff points must be increased. When choosing VG1 = 12V, the cutoff points will be adjusted to 170V (instead of 150V). Since the power supply is 200V, 30V are available to ensure correct blanking operation. The DC output voltage must be increased by 12V from its previous value. Note that all the phenomena described in this section concern a static or quasi-static (15kHz) operation (e.g. white picture or rather large white pattern on a black background). When current peaks occur (e.g white characters insertion or straight luminance transition), the peaks will be absorbed by the coupling capacitor and the voltage amplifier,and hence the tube will be able to source a greater current. IV.5 - Low Current Measurements We have seen in section II-2.2 how the beam current monitoring works (see Figure 3). We have
13/21
Another solution consists of connecting a zener diode as shown in Figure 15.With this device the
5101A-17.EPS
= 50 V
R eq
TEA5101A APPLICATION NOTE
seen that the capacitor C must charge again after the blanking phase. This charge is generally furnished by the tube capacitor independently from the beam current.However,if during the blanking phase, the output voltage is too low (e.g. the PMOS is reverse biased (-20V) because of a too high leakage current or when measuring with an oscilloscope probe), the VC required to charge C again will be greater than the maximum charge available from the tube capacitor.Hence the beam current will have to charge C in a first step. Since this current is rather low during the cutoff adjustement phase, a long time will be spent to charge C. The current absorbed by the PMOS and fed to the videoprocessor will not be equal to the beam current and the cutoff adjustementwill not be correct. Hence the reverse voltage across the capacitor C must be limited by a diode connected as follows : Figure 16
1N4148
to be connected. Both beam current monitoring modes (sequential and parallel) are possible. The layout and the electrical diagram of the PC board are shown in Figures 19 and 20. V.2 - Performance Evaluation As seen in chapter IV, the dynamic performances (bandwidth, crosstalk) of the TEA5101A is very dependent on the PCB layout.Consequently, the evaluation board has been designed to obtain the best results. To evaluate the performance, the best way is to work outside of the TV set by driving the amplifier by an HF generator (or a network analyser) while simulating the load conditions fixed by the CRT, since AC performances are directly determined by the load. V.2.1 - Measurement conditions The schematic diagrams of the AC measurements are shown in Figures 21 and 22. The conditions are as follows : - B I AS I NG : VOU T D C = 10 0V by choosing R11 = R21 = R31 = 1.5k and VDD = 200V - AC GAIN = 50 by adjusting P10, P20, P30 - LOADING : - by a 8.2pF capacitor and the probe capacitor (2pF), the sum is equivalent to the capacitance of a CRT with the socket and the spark gaps - the 1M resistors connected between each output and VDD allow the conduction of the beam current monitoring PMOS transistor in such a way that VADC = VBDC= 100V. - DRIVING by a 1F capacitor, the HF generator being loaded by 50. - the dynamic power dissipated in the IC will increase with frequency. To avoid the temperature increasing, it is necessary to do very quick measurements or to use a low Rth (7oC/W) heatsink in forced convection configuration. Such conditions are not present in a TV set since the driving signal will be a video signal instead of a pure HF signal. V.2.2 - Results V.2.2.1 - Bandwidth The curves Figures 23 and 24 show the frequency responses of one channel with 100Vpp and 50Vpp output voltages. The bandwidths are approximatively 8MHz at 100Vpp and up to 10MHz at 50Vpp. V.2.2.2 - Crosstalk The curves Figures 25, 26 and 27 show the crosstalk for this application.Thecrosstalk is almost the same for the six different combinations of the three channels. The worst value is -24dB at 5MHz.
C
7
5101A-18.EPS
9 6
With this configuration, the voltage across C will be -0.6V max. Since this voltage must be 2.5V in the stationary state (see section II-2.2), the voltage across C must be increased by 3.1V and this charge can be supplied by CL. We can also slightly decrease the value of C. However if C is too low, the HF behaviour will be impaired. V - APPLICATION EXAMPLES V.1 - Application Description Figures 17 and 18 show two applications, one for a 45AX tube and the videoprocessor TDA3562A (application 1), the other designed for S4 type tube and the videoprocessor TEA5031D (application 2). In these two applications, the nominal gain is 28dB and the output black level is 150V. The quasi cutoff currents are respectively 10A and 16A for applications 1 and 2. These applications are implemented using the same PC board especially designed to allow different options for tube biasing, power supply decoupling and connections. This PC board allows also two different tube sockets (jedec B8274 or B10277)
14/21
TEA5101A APPLICATION NOTE
Figure 17 : Application 1 (45AX Tube, TDA3562A) - Electrical Diagram
TEA5101A
R12 39k 1W P10 2.2k R in R11 820 R10 390
4
9
VDD
C10 470pF R13
7
R CATHODE 1k 1/2W
V REF
6
R14 4.7k
R22 39k 1W P20 2.2k G in R21 820 R20 390
3
12
VDD
C20 470pF R23
10
12V G CATHODE 120k CHASSIS 68k B CATHODE VDD 200V CRT GROUND
1k 1/2W V REF
11
R24 4.7k
R32 39k 1W P30 2.2k B in R31 820 R30 390
1
15
VDD
C30 470pF R33
13
1k 1/2W V REF
2 8 5
14
R34 4.7k 47 C2 10mF 250V C1 0.1F 250V G1
VCC 12V CHASSIS GROUND
C4 10F 63V
C3 0.1F
HEATER I/O LOW LEVEL CONNECTOR C1 I/O HIGH LEVEL CONNECTOR C2 CRT CONNECTOR PIN CONNECTION G2 1nF 630V
5101A-19.EPS
10k 1/2W
15/21
TEA5101A APPLICATION NOTE
Figure 18 : Application 2 (PIL24 Tube, TEA5031D) - Electrical Diagram
TEA5101A
R12 39k 1W P10 2.2k R in R11 820 R10 390
4
9
VDD
C10 1nF R13
7
R CATHODE 1k 1/2W R CUT-OFF R14 4.7k R15 100k
V REF
6
R22 39k 1W P20 2.2k G in R21 820 R20 390
3
12
VDD
C20 1nF R23
10
G CATHODE 1k 1/2W G CUT-OFF R24 4.7k R25 100k
V REF
11
R32 39k 1W P30 2.2k B in R31 820W R30 390
1
15
VDD
C30 1nF R33
13
B CATHODE 1k 1/2W B CUT-OFF R34 4.7k R35 100k
V REF
2 8 5
14
V CC 12V CHASSIS GROUND
47 C4 10F 63V C3 0.1F C2 10F 250V C1 0.1F 250V G1
VDD 200V
CRT GROUND
HEATER R1 100k I/O LOW LEVEL CONNECTOR C1 I/O HIGH LEVEL CONNECTOR C2 CRT CONNECTOR PIN CONNECTION G2 2 x 22nF 630V
10k 1/2W
5101A-20.EPS
16/21
TEA5101A APPLICATION NOTE
Figure 19 : TEA5101A Evaluation Board Layout and Components View COMPONENT SIDE
COPPER SIDE
17/21
5101-21A.EPS / 5101-21B.TIF
TEA5101A APPLICATION NOTE
Figure 20 : TEA5101A Evaluation Board Electrical Schematic Diagram
TEA5101A
R12 39k 1W P10 2.2k R in R11 1.5k R10 390
4
9
VDD
C10 1nF R13
7
R CATHODE 1k 1/2W R CUT-OFF R14 4.7k R15 120k
V REF
6
R22 39k 1W P20 2.2k G in R21 1.5k R20 390
3
12
VDD
C20 1nF R23
10
G CATHODE 1k 1/2W G CUT-OFF R24 4.7k R25 120k
V REF
11
R32 39k 1W P30 2.2k B in R31 1.5k R30 390
1
15
VDD
C30 1nF R33
13
B CATHODE 1k 1/2W B CUT-OFF R34 4.7k R35 120k
V REF
2 8 5
14
47 V CC 12V CHASSIS GROUND C4 10F 63V C3 0.1F C2 10F 250V C1 0.1F 250V G1 VDD 200V
CRT GROUND
HEATER R1 100k I/O LOW LEVEL CONNECTOR C1 I/O HIGH LEVEL CONNECTOR C2 CRT CONNECTOR PIN CONNECTION G2 2 x 22nF 630V 10k 1/2W
5101A-22.EPS
18/21
TEA5101A APPLICATION NOTE
Figure 21 : Bandwidth Measurement Configuration
VDD VDD
P10 1F 2.2k
R10 390
R12 39k 1W A
C 10 1nF
1M
R13
100nF
1k 1/2W
50k
R11 1.5k
B
V REF
8P2
ACTIVE PROBE -40dB CL 2pF
PASSIVE PROBE -20dB
S HP 3577 NETWORK ANALYSER
R ATTENUATOR -20dB
5101A-23.EPS
A
Figure 22 : Crosstalk Measurement Configuration
VD D V DD
4.7k
P10 1F 2.2k
R10 390
R12 39k 1W
A
C10 1nF B V REF
1M
R13 VD D
100nF
1k 1/2W
8P2
50k
R11 1.5k
ACTIVE PROBE -40dB CL 2pF
S HP 3577 NETWORK ANALYSER
R A ACTIVE PROBE -40dB
ATTENUATOR -20dB
V DD
4.7k
P20 1F 2.2k
R20 390
R32 39k 1W
A
C20 1nF
1M
R23
100nF
1k 1/2W
R21 1.5k
B V REF
8P2
4.7k
19/21
5101A-24.EPS
TEA5101A APPLICATION NOTE
Figure 23 : Frequency Response of R Channel (100VPP) Figure 24 : Frequency Response of R Channel (50VPP)
5101A-25.TIF
Figure 25 : Crosstalk between R Channel and G and B Ones
Figure 26 : Crosstalk betweenGR Channel and R and B Ones
5101A-27.TIF
Figure 27 : Crosstalk between B Channel and R and G Ones
20/21
5101A-29.TIF
5101A-28.TIF
5101A-26.TIF
TEA5101A APPLICATION NOTE
V.2.2.3 - Transition times The curves Figure 28 show respectively the R, G, B rise and fall times of respectively 49ns and 48ns with a 100Vpp output voltage (between 50 and 150V). The difference between rise times of the three channels is less than 1ns. The difference between fall times of the three channels is less than 2ns. The delay time at rising output is 48ns. The delay time at falling output is 50ns. The differencebetween the delay times is less than 2ns. The slew rate is about 2000V/s. These results verify the high performance available from the TEA 5101A video amplifier which make it very suitable for high end applications.
Figures 28A and 28B : TEA5101A R Channel Step Response
Information furnished is believed to be accurate and reliable. However, SGS-THOMSON Microelectronics assumes no responsibility for the consequences of use of such information nor for any infringement of patents or other rights of third parties which may result from its use. No licence is granted by implication or otherwise under any patent or patent rights of SGS-THOMSON Microelectronics. Specifications mentioned in this publication are subject to change without noti ce. This publication supersedes and replaces all information previously supplied. SGS-THOMSON Microelectronics products are not authorized for use as critical components in life support devices or systems without express written approval of SGS-THOMSON Microelectronics. (c) 1994 SGS-THOMSON Microelectronics - All Rights Reserved Purchase of I2C Components of SGS-THOMSON Microelectronics, conveys a license under the Philips I2C Patent. Rights to use these components in a I2C system, is granted provided that the system confo rms to the I2C Standard Specifications as defined by Philips. SGS-THOMSON Microelectronics GROUP OF COMPANIES Australia - Brazil - China - France - Germany - Hong Kong - Italy - Japan - Korea - Malaysia - Malta - Morocco The Netherlands - Singapore - Spain - Sweden - Switzerland - Taiwan - Thailand - United Kingdom - U.S.A.
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5101A-31.TIF
5101A-30.TIF

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